6418213.pdf

design ideas

EditEd By Brad thompson

and Fran GranvillE

readerS SOLVe deSIGN PrOBLeMS

Current mirror improves

PWM regulator’s performance

D Is Inside

70 Low-cost current monitor

tracks high dc currents

74 Digital-I/O circuit adapts to

many interface voltages

E What are your design problems

and solutions? Publish them here

and receive $150! Send your

Design Ideas to edndesignideas@

reedbusiness.com.

Power-supply designs requiring

high-performance isolated feed-

back often use an error amplifier simi-

lar to the one in Figure 1 , which relies

on a second amplifier, IC 1B , to provide

the necessary inversion to keep the op-

tocoupler, IC 2 , referenced to ground.

To prevent bias-supply noise from en-

tering the feedback path and causing

oscillations, the amplifier relies on its

ground reference and power-supply-

rejection characteristics. The power

supply’s output drives a voltage divider

comprising R 1 and R 2 that maintains

the amplifier’s inverting input at the

same voltage as the reference voltage

that IC 3 provides. C 2 , R 3 , and C 3 com-

prise frequency-compensation com-

ponents for the power supply’s stable

operation. This component-intensive

error-amplifier configuration requires

two operational amplifiers, one preci-

sion shunt-voltage reference, four ca-

pacitors and often a fifth in parallel

with R 6 , and seven resistors.

Figure 2 shows an alternative sin-

gle-amplifier design in which IC 3 , an

LM4040 precision-voltage reference,

drives optocoupler IC 2 with a “stiff”

positive-voltage source over a wide

current range. The voltage reference

suppresses any noise present on the

bias-supply rail. Variations in the ref-

erence and power-supply voltages ap-

pear in common mode at the amplifi-

er’s inputs and thus provide addition-

al noise immunity. A resistive-voltage

divider comprising R 2 and R 3 reduc-

es the reference voltage to equal the

power supply’s regulated output volt-

age, which drives IC 1 ’s inverting input

through R 1 . Given its single voltage di-

vider, the error-amplifier circuit of Fig-

ure 2 provides the same output voltage

as the circuit of Figure 1 and requires

a single operational amplifier and pre-

cision shunt reference, four capacitors,

and six resistors.

Miller-effect coupling of collector-

emitter-voltage transitions into a typi-

cal phototransistor-based optocoupler’s

high-impedance, optically sensitive

V CC

R 4

C 2

IC 3

LM4040

2.5V

VOLTAGE

REFERENCE

R 3

C 3

R 6

FROM

POWER

SUPPLY’S

OUTPUT

C 1

0.1 �F

R 1

�

R 5

�

IC 1A

R 7

IC 2

IC 1B

R 2

TO REMAINDER

OF PWM POWER

SUPPLY

C 4

Figure 1 A conventional isolated-feedback circuit requires an extra operational amplifier and adds several passive compo-

nents to a representative pulse-width-modulated power-supply design.

march 1, 2007 | EDN 69

Grant smith, national semiconductor, phoenix, aZ

design ideas

base region introduces a bandwidth-

limiting pole, which dramatically slows

the device’s response time. Holding

the phototransistor’s collector-emit-

ter voltage constant and allowing only

its collector-emitter current to change

provide an order-of-magnitude switch-

ing-speed improvement.

National Semiconductor’s (www.

national.com) LM5026 active-clamp

current-mode PWM controller, IC 4 ,

provides a convenient method of re-

ducing an optocoupler’s Miller-effect-

induced slowdown. Figure 2 shows the

LM5026’s internal current mirror driv-

ing what would normally serve as a fre-

quency-compensation pin. Optocou-

pler IC 2 connects directly between two

constant-voltage sources comprising

the current mirror and a voltage ref-

erence. The resultant decrease in re-

sponse time relocates the bandwidth-

limiting pole and improves the circuit’s

transient response.

The values of C 2 , C 3 , R 3 , and R 1 apply

only to this design and may require

modification for other applications.

Select R 1 to provide equal impedances

at both of the op amp’s inputs. C 2 forms

a high-frequency noise filter. After you

measure the converter’s overall gain,

calculate values for C 3 and R 3 that

will provide proper gain and phase re-

sponse. Several methods of calculation

are available, most of which will pro-

vide adequate results. EDN

V CC

R 5

499

IC 4

LM5026

R 6

C 2

100 pF

IC 3

LM4040

4.1V

5V REF

R 4

10k

C 3

0.033 �F

C 4

1000 pF

IC 2

R 2

30.9k

REF

C 1

0.1 �F

TO REMAINDER

OF PWM POWER

SUPPLY

FROM

POWER SUPPLY’S

OUTPUT

COMP

�

IC 1

LM8261

R 1

24.9k

PWM

COMPARATOR

R 3

127k

CURRENT

MIRROR

Figure 2 Clamping an optoisolator’s voltage excursion improves the PWM-regulator loop’s transient response.

Low-cost current

monitor tracks

high dc currents

susanne nell, Breitenfurt, austria

CORE REACHES

SATURATION

0.3

B(H)

MAGNETIC-

FLUX DENSITY

(TESLA)

0.2

PERMEABILITY

To measure high levels of direct

current for overload detection

and protection, designers frequently

use either a current-shunt resistor or a

toroidal core and Hall-effect magnetic-

field sensor. Both methods suffer from

drawbacks. For example, measuring

20A with a 10-m V resistor dissipates

4W of power as waste heat. The Hall-

effect sensor delivers accurate measure-

ments and wastes little power, but it’s

0.1

MAGNETIC-FIELD STRENGTH

(A/m)

Figure 1 This representative magnetization (BH) curve shows that, as current

through an inductor’s winding increases, so does magnetizing-field strength,

H. When magnetic-flux density, B, can no longer increase, the core’s magnetic

material has reached saturation.

70 EDN | march 1, 2007



design ideas

an expensive approach to simple cur-

rent monitoring.

This Design Idea describes an inex-

pensive, low-power current-measure-

ment circuit that’s useful for measure-

ments of modest accuracy. As a bonus,

a filter inductor in a dc/dc converter’s

input line can double as a current sen-

sor for the measurement circuit. A rep-

resentative ferrite core’s permeability

decreases as the core nears saturation

( Figure 1 ). The curve’s shape and val-

ues depend on the core material’s char-

acteristics and whether the core in-

cludes an air gap.

The core’s permeability depends on

the magnetic-flux level in the ferrite

material, which in turn depends on

the amount of current flowing through

the core’s windings. This circuit uses

a simple LC oscillator to measure the

core’s permeability. A primary winding

J 4

CURREN T TO MEASURE

I DC

J 3

OUT

PRIMARY: ONE TURN

T 1

SECONDARY: 50 TURNS

TOROID FERRITE CORE

J 2

R 2

47k

R 3

6.8k

C 4

100 nF

FREQUENCY

OUTPUT

POWER INPUT

Q 1

BC548

C 2

22 nF

J 1

C 1

10 �F

10V

V CC

GND

R 1

15k

R 4

2.2k

C 3

47 nF

Figure 2 Varying the direct current flowing through the single-turn primary

winding alters T 1 ’s secondary winding’s inductance, which in turn varies the

oscillator’s output frequency.

a fILter INductOr

IN a dc/dc cON

Verter’S INPut LINe

caN dOuBLe aS a

curreNt SeNSOr fOr

the MeaSureMeNt

cIrcuIt.

OUTPUT

FREQUENCY

(kHz)

SIEMENS

FONOX

VAKUUMSCHMELZE

comprising one or more turns wound

on the core carries the measurement

current. A multiturn secondary wind-

ing on the core forms an inductor, L,

that determines the oscillator’s reso-

nant frequency.

In theory, any LC oscillator circuit

will serve in this application, but, in

practice, the current-measurement

winding presents a low impedance that

damps the LC-tank circuit and causes

start-up and stability problems in some

oscillator circuits. Of a variety of test-

ed oscillator circuits, the design in Fig-

ure 2 offers the best performance. A

number of factors affect the core’s per-

meability, which in turn impacts the

circuit’s frequency stability and lim-

its its applications to current-overload

detection and low-accuracy current

measurements.

Figure 3 illustrates the circuit’s out-

9 10 11

DC PRIMARY CURRENT (A)

Figure 3 Current-versus-output-frequency plots for three manufacturers’ toroi-

dal cores show the influence of the cores’ characteristics on frequency linear-

ity and relative sensitivity.

put-frequency-versus-current charac-

teristics for three vendors’ ferrite cores

of identical dimensions and number of

secondary turns. For best linearity, use

a low-hysteresis core material. Cores of

virtually any dimensions and materials

work in the circuit but require optimi-

zation of the number of turns on the

oscillator tank and primary windings.

Increase the core’s air gap, if present,

when the current you apply to the core

causes saturation before reaching the

overload value. For improved perfor-

mance and linear measurements, use

the circuit in a closed-loop configura-

tion ( Reference 1 ). EDN

R e f e R e n c e

Nell, Susanne, “Improved current

monitor delivers proportional-voltage

output,” EDN , Jan 19, 2006, pg 84,

www.edn.com/article/CA6298271.

72 EDN | march 1, 2007

design ideas

Digital-I/O circuit adapts

to many interface voltages

steve hageman, Windsor, Ca

To solve the problem, I designed a

flexible digital-interface circuit around

a MAX7301 I/O expander from Max-

im Integrated Products (www.maxim-

ic.com) and a programmable linear-

power supply comprising a MAX1658

adjustable linear-voltage regulator un-

der the control of a MAX5400 256-

position, digitally programmable po-

tentiometer. This circuit provides a

programmable interface matching the

logic levels of ICs that require 2.5, 3,

3.3, and 5V power supplies.

Two SPIs (serial-peripheral inter-

To test products in my R&D

lab, I build many universal data-

acquisition systems that connect to a

PC or another controller through RS-

232 links or LANs. These small sys-

tems typically include multiple ADC,

DAC, and digital-I/O channels to con-

trol various hardware functions during

product design and development. Over

the years, I have established a simpli-

fied analog-interface standard that

spans a 0 to 5V range. On the digital

side, many of the newer logic families

no longer tolerate 5V inputs and have

rendered 5V-only digital-I/O ports ob-

solescent.

ADJUSTABLE REGULATOR

DIGITAL-I/O VOLTAGE

TO Q 1 TO Q 6

2.5 TO 5V

5.25V

IC 3

MAX1658

GND

VOUT

C 1

10 �F

C 2

0.1 �F

VIN

C 3

0.1 �F

C 4

10 �F

SET

5.25V

R 7

86.6k

R 1

4.7k

VCC

Q 1

SI1012R/X

SCLK

DIN

IC 2

MAX5400

GND

256-POSITION,

50-k� TRIM

POTENTIOMETER

C 5

0.1 �F

CHIP SELECT

MAX5400

R 8

41.2k

5.25V

DIGITAL-I/O VOLTAGE

R 2

4.7k

R 4

4.7k

SCLK

DIN

DOUT

VCC

P12

P13

P14

P15

P16

P17

P18

P19

P20

P21

P22

P23

P24

P25

P26

P27

P28

P29

P30

P31

Q 4

SI1012R/X

Q 2

SI1012R/X

SCLK

ISET

5.25V

DIGITAL-I/O

VOLTAGE

R 9

39k

20 PROGRAMMABLE

DIGITAL-I/O LINES

WITH VARIABLE

VOLTAGE SPANNING

2.5 TO 5V

IC 1

MAX7301AAI

R 3

4.7k

R 5

4.7k

Q 3

SI1012R/X

Q 5

SI1012R/X

DATA

MICROPROCESSOR

CONTROLLER

1.8 TO 5V LOGIC

DIGITAL-I/O

VOLTAGE

CHIP SELECT

MAX7301

R 6

4.7k

GND

SCLK

DATA

Q 6

SI1012R/X

Figure 1 A programmable power supply sets voltage thresholds for a universal digital-I/O device.

74 EDN | march 1, 2007



design ideas

faces) control all 20 of the MAX-

7301AAI’s input and output pins and

voltage thresholds ( Figure 1 ). Un-

like some SPI-port expanders that in-

clude weak, resistor-only pullups, the

MAX7301, IC 1 , features true, active-

pullup, “totem-pole” outputs that can

source higher currents. When powered

by the SPI-programmable linear regu-

lator, the MAX7301’s outputs can de-

liver logic levels of 2.5 to 5V. The pro-

gramming interfaces for both devices

comprise two three-wire (plus ground)

SPI connections that use only six of

the controller’s signal lines.

Six Vishay (www.vishay.com) Si-

1012R low-gate-voltage-threshold N-

channel MOSFETs, Q 1 through Q 6 ,

isolate the controllers’ fixed-output-

voltage levels from IC 1 ’s variable-in-

put-threshold voltages. Although any

of several IC-level-translator ICs work

equally well, the inexpensive MOSFET

buffers occupy small footprints on the

interface’s PCB (printed-circuit board).

For operation at serial-interface clock

uNLIke SOMe SPIPOrt

exPaNderS that

INcLude weak, reSIS

tOrONLy PuLLuPS, the

Max7301, Ic 1 , featureS

true actIVePuLLuP,

“tOteMPOLe” OutPutS

that caN SOurce

hIGher curreNtS.

trols IC 3 , a Maxim MAX1658 adjust-

able-voltage linear regulator. Writing

all zeros to IC 2 sets IC 3 ’s output voltage

to slightly more than 5V, and writing

all ones (255 decimal) to IC 2 reduc-

es IC 3 ’s output voltage to slightly less

than 2.5V. To compensate for compo-

nent tolerances, the circuit provides

enough voltage overrange to cover the

full 2.5 to 5V range. Writing 128 (dec-

imal) to IC 2 should produce a nomi-

nal 3.25V output. Measure IC 3 ’s actual

output voltage and subtract it from the

nominal voltage to produce an offset

count for calibration correction.

In operation, the host controller sets

IC 3 ’s regulated output voltage through

IC 2 and determines the maximum volt-

ages of IC 1 ’s logic inputs and outputs.

Next, the controller configures IC 1 ’s in-

puts and outputs as necessary for the in-

terface task at hand. The MAX7301’s

standard CMOS logic-threshold voltag-

es of 0.3 to 0.7 times its supply voltage

for low and high inputs, respectively, in-

terface with other CMOS parts. EDN

rates approaching IC 1 ’s 26-MHz maxi-

mum, optimize the values of resistors

R 1 through R 6 to provide adequate rise

times at the selected clock rate. These

values are adequate for operation at the

1-MHz SPI clock rate that a low-power

microcontroller produces.

To alter the circuit’s output-volt-

age level, IC 2 , a 256-step Maxim

MAX5400 digital potentiometer, con-

76 EDN | march 1, 2007

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